This invention relates to the field of DC-to-DC power conversion and in particular to the class of converters distinguished by ultra high efficiency, high overload capability, small size and weight, high power density at a moderate switching frequency.
The following notation is consistently used throughout this text in order to facilitate easier delineation between various quantities:
1. DCxe2x80x94shorthand notation historically referring to Direct Current but now has wider meaning and refers to all Direct electrical quantities (current and voltage);
2. ACxe2x80x94shorthand notation historically referring to Alternating Current but now has wider meaning and refers to all Alternating electrical quantities (current and voltage)
3. The instantaneous time domain quantities are marked with lower case letters, such as i1 and v2 for current and voltage. Often these instantaneous quantities contain a DC component, which is designated with corresponding capital letters, such as I1 and V2.
4. The difference between instantaneous and DC components is designated with xcex94, hence xcex94i1 designates the ripple component or AC component of current i1.
Over the last two decades a large number of switching DC-to-DC converters had been invented with the main objective to improve conversion efficiency and reduce the converter size. The past attempts to meet both of these objectives simultaneously have been hampered by the two main obstacles, which up to now seemed to be inherent to all switching DC-to-DC converters:
1. The large DC current bias present in the filtering inductors at either input or output of the converters (as well as the DC-bias current present in the isolation transformer of some of the isolated converters) resulted in a big size of the magnetic components, since an air-gap proportional to the DC current bias must be inserted in the AC flux path in order to prevent magnetic core saturation. This also resulted in a very inefficient use of the magnetic material, which was largely wasted. Even a relatively small air-gap, in the order of 1 mm (40 mils), drastically reduces the total inductance. This loss of inductance was compensated by either an inordinately large increase of the switching frequency (hence increase of losses) or by increasing the size of the magnetic cores, or both.
2. An implementation of soft switching methods to reduce significant switching losses at increased switching frequencies was DC load current dependent and required for its operation an unwanted large output inductor AC current ripple (larger than twice the magnitude of the maximum DC load current) thereby diminishing most of the recovered energy due to increased conduction losses caused by this large AC ripple current.
Magnetic Saturation with DC Current Bias
First, the problem associated with the DC-bias of magnetic components (inductors and transformers) can be best understood with reference to the classical buck converter shown in prior art FIG. 1a and the accompanied output inductor current waveform of FIG. 1b. Since the converter output supplies DC power to the load, the inductor in the buck converter must pass the DC component of the load current, which is IDC. Hence, it clearly cannot be designed as an ordinary inductor used in alternating current (AC) applications such as the inductor in FIG. 2a. 
Several quantities which are used throughout the text are now described with their defining relationship:
Flux linkage xcex is the total flux linking all N turns and is xcex=N"PHgr" where "PHgr" is the flux in the magnetic core;
Flux density B is the flux per unit area defined by B="PHgr"/S where S is a magnetic core cross-section area.
Inductance L is defined as the slope of xcexxe2x88x92i characteristic, i.e., L=xcex/i;
Duty ratio D of the switch is defined as D=tON/TS where tON is ON time of the switch, and TS is the switching period defined as TS=1/fS where fS is a constant switching frequency;
Complementary duty ratio Dxe2x80x2 of the switch is defined as Dxe2x80x2=1xe2x88x92D.
An AC inductor is wound on magnetic core material in order to dramatically increase its inductance value. For example, typical ferrite core material has at room temperature a relative permeability on the order of xcexcr.3,000. Hence the inductance of the coil is magnified by a factor of 3,000 simply by inserting the magnetic core material without any air gap as in FIG. 2a. The corresponding flux linkage xe2x80x9cxcexxe2x80x9d versus current xe2x80x9cixe2x80x9d characteristic is as in FIG. 2b with a high slope illustrating the high inductance value L (maximum attainable with that core material). The flux linkage excursions (caused by the AC current) are symmetrical around the center of the magnetic core operating characteristic. Even if a very small DC current IDC shown in FIG. 2b were to pass through this coil, the magnetic core material would saturate and instead of the desirable large inductive impedance, the inductor would look like a short circuit. Thus, to avoid core saturation, all present switching converters xe2x80x9csolvexe2x80x9d this DC-bias problem in a xe2x80x9cbrute-forcexe2x80x9d way by inserting an air-gap in the magnetic flux path as illustrated in FIG. 3a. This clearly reduces the inductance value proportionally to the inserted air-gap size (the larger the DC current, the bigger air-gap is needed, hence the smaller is the resulting inductance value), as seen by the flux linkage characteristic of FIG. 3b for an un-gapped and gapped core and their corresponding inductances L and Lg. Clearly three very detrimental factors did occur:
1. By insertion of the air-gap, the inductance value is drastically reduced. It is not uncommon to see the original un-gapped inductance L reduced by a factor of 100 to 1000 to the inductance Lg with the air-gap included. In order to compensate for this loss of inductance, the switching frequency is radically increased or a much bigger core size is used, or both.
2. The already small AC flux linkage excursions due to the finite and relatively low saturation flux density BSAT of 0.3 T (tesla) for ferrite material, is further significantly reduced due to the presence of the DC-bias in the core. For example, in typical applications, the DC-bias might correspond to a flux density of 0.25 T thus leaving only 0.05 T for the superimposed AC flux excursions. This in turn results in either larger core size requirements or increased switching frequency, or both.
3. The waste of ferromagnetic material is even larger, since the negative part of the saturation characteristic is not utilized at all, and thus another xcex94B=BSAT=0.3 T is also wasted.
The DC-bias problem is not only limited to all inductors used up to now in DC-to-DC converters but is also present in many isolation transformers, such as for example in the popular flyback converter shown in FIG. 4a. This transformer does provide galvanic isolation and the ability to step-up or step-down the voltage through the transformer turns ratio, but contrary to the ordinary AC line transformer, it has a large DC-bias and requires a correspondingly big air-gap as shown in FIG. 4b. Hence the magnetic core is biased in one direction thus limiting the superimposed AC flux excursions as seen in FIG. 5.
Up to now, the detrimental effect of the large DC-bias and hence the large air-gap was introduced qualitatively. Let us now also quantify these effects on an output inductor design example for a 5V, 100 W buck-like converter, having a DC load current I2=20 A, and number of winding turns N=6, which is implemented on a ferrite core having a saturation flux density BSAT=0.3 T (tesla) out of which BDC=0.2 T is available for the DC-bias and the remaining 0.1 T is allocated for the superimposed AC flux excursions. To support NI=120 ampere-turns the required air-gap is given by formula lg=xcexc0NI/BDC=30 mils=0.75 mm where xcexc0=4xcfx8010xe2x88x927 H/m is the permeability of free space. If L is the inductance without air-gap, and Lg is the inductance with air-gap Lg=30 mils, then the ratio of the two inductances is given by L/Lg=xcexcrlg/lm=50 where xcexcr is the relative permeability of the ferrite material, which for typical materials used in switching converters is xcexcr=3000, and lm=4.5 cm is the magnetic path length of the core used. Thus, even at a relatively modest DC current level of only 20 A, the maximum available inductance of a given core is reduced by a factor of 50. The factor by how much presence of ferrite increases the coil inductance above that of an air-core is xcexcr which is 3000 in our example. Then, the ratio of 3000/50=60 is how much the addition of the ferromagnetic material with air-gap increases the inductance of the inductor built with no ferromagnetic material but with the same cross-section. At higher power and especially DC load current levels this becomes progressively much more severe. It is not uncommon for some high power DC converter applications in the kilowatt range to see that after ferromagnetic material was inserted, the inductance increased only by a factor of 2 or 3 over the inductance without any magnetic material due to the large air-gap needed to prevent saturation. Clearly, this is a tremendous waste of the magnetic material which has the ability to increase the inductance 3000 times over that of an air-core coil. This is also the reason, why in switching converters in which isolation transformer has no DC bias, such as in the isolated Cuk converter, the transformer size is several times smaller in size and weight in comparison with the size and weight needed for the input and output inductors, which by far dominate the size and weight of the switching converter and also result in increased losses.
In the above typical example, the loss of the inductance due to insertion of the air-gap in the flux path is compensated either by increasing the core cross-section thus making the converter size substantially bigger, or by increasing the switching frequency by an order of magnitude, or a combination of both. This clearly would rapidly degrade the overall efficiency, even in the presence of soft-switching. Thus, it would be very desirable to either substantially reduce the DC-bias in the magnetic core, or, if possible, to eliminate it entirely.
In the past, there had been a number of attempts to correct this fundamental limitation of DC-to-DC converters, but with a very limited success. One approach was followed by magnetic manufacturers, such as Hitachi and TDK. In the article xe2x80x9cReducing Magnetic Component Size with Reverse Biased Ferrite Corexe2x80x9d published in the Proceedings of the Powercon 6 conference, May 1979, author Shiraki (of Hitachi) proposed to add a permanent magnet to the air-gap and hence by proper orientation of the permanent magnet reverse bias the core in the direction opposite to the DC-bias created by the current of the magnetics winding as shown in FIG. 6a. The net effect is that the AC flux excursions are now extended into the negative core flux swing area as seen in FIG. 6c and would allow the core cross-section and volume reduction by up to 50%. The TDK corporation developed a line of PCH cores based on their reverse biased core modification as reported in the Proceedings of Powercon 9, July 1982 in article xe2x80x9cA New Reverse Biased Choke Coilxe2x80x9d by Nakamura and Ohta of TDK corporation. Note, however, that both approaches are operated with additional air-gap, that is along the reduced, xe2x80x9cheavy linexe2x80x9d slope as shown in FIG. 6b and FIG. 6c. Hence, the large inductance reduction from its maximum inductance capability of the un-gapped core (dashed line in FIG. 6b and FIG. 6c) is still present. Clearly, the core can only support the designed-in maximum DC-ampere-turns. If that is exceeded, the core will saturate and the overload capability will be lost. Since the permanent magnet provides a fixed reverse bias independent of the DC load current, at no-load current, the core flux is entirely along the negative part of the core flux saturation characteristic (FIG. 6c). In fact, the permanent magnet generates the maximum allowable bias but in the negative (reverse) direction. This will be compared later with the novel DC Transformer switching converter in which there is an automatic self-balancing, such that at any DC load current there is no net DC-bias and no DC flux in the core.
In addition to the above limited performance improvements, the other practical limitations, such as increased cost of the special cores with inserted permanent magnets, the extra loss due to added core loss of the permanent magnet, etc., rendered this approach unattractive, which is by now abandoned by both of these companies.
Another attempt was made to use a special converter circuit configuration instead of a special magnetic core structure to reduce or eliminate the DC-bias problem. Such an approach is disclosed in U.S. Pat. No. 5,166,869 issued to Bryce L. Hesterman for xe2x80x9cComplementary Electronic Power Converterxe2x80x9d in which a xe2x80x9ccomplementary transformerxe2x80x9d is introduced. This transformer combines the input and output inductors into a coupled-inductor configuration in which the DC flux generated by the input inductor DC current is canceled by the flux generated by the output inductor DC current. The main drawback of the proposed converter is that it is capable of producing only the fixed input to output voltage conversion ratio determined by a fixed turns ratio of the two windings. Hence it cannot provide a regulated voltage through pulse-width modulation of the switches even over a limited input voltage range. From another point of view, there are other fixed conversion ratio converters such as 50% driven bridge type converters, which do not need inductors with DC-bias current for either input or output filtering, hence the DC-bias problem is not present.
Thus, a highly desirable objective is to have a switching converter with a variable conversion ratio, capable of handling a wide range of input voltages and provide regulated output, and at the same time either completely eliminate the DC-bias or reduce it substantially.
Another possible approach is to combine input and output inductor windings into a common coupled-inductor structure as shown in FIG. 7a and as was disclosed in U.S. Pat. No. 4,184,197, xe2x80x9cDC-to-DC Switching Converterxe2x80x9d by S. Cuk and R. D. Middlebrook and U.S. Pat. No. 4,186,437, xe2x80x9cDC-to-DC Switching Converter with Zero Input and Output Current Ripple and Integrated Magnetics Circuitsxe2x80x9d by S. Cuk. As described in the above patents, the basic prerequisite for combining the two windings on a common core is to have identical AC voltages across the two inductors before the coupling, and that the AC voltage matching is maintained over a wide operating range of duty ratio D as illustrated by the identical AC voltage waveforms in FIG. 7b (duty ratios D1 and D2) for the converter of FIG. 7a. In practical applications, a small mismatch of the AC voltages could be absorbed gracefully due to the ever-present leakage inductance between the two windings as explained below.
Since the AC voltages are identical, the placement of the two windings on the same core in a coupled-inductor structure imposes the requirement for equal number of turns N (AC voltage ratio equal to turns ratio as in an ideal transformer), because in the simplified model the leakage inductance is not included. The proper understanding of the AC voltage polarity marking in coupled-inductor and integrated magnetic structures (polarity markings with dot-marked ends as in FIG. 7a) and the actual directions of the instantaneous and DC currents relative to these dot markings (currents i1 and i2 and their DC components I1 and I2 in FIG. 7a) is of critical importance for understanding not only previous inventions but is crucial for understanding the present invention.
Note the difference of this coupled-inductor structure and a transformer. The output inductor instantaneous current i2 in the coupled-inductors of FIG. 7a flows into the dot-marked end, whereas in an AC transformer, the secondary current i2 flows out of the dot-marked terminal. Clearly, the air-gaps g1 and g2 of the two corresponding separate inductors of FIG. 8a and FIG. 8b add, resulting in larger total air-gap g1+g2 for the coupled-inductor core structure of FIG. 9a. Thus, the corresponding DC component I2 of the load current in the coupled-inductor structure also flows into the dot-marked end. Consequently, the generated DC fluxes "PHgr"1 and "PHgr"2 add together (FIG. 9a) resulting in a combined flux vs. ampere-turns characteristic of FIG. 9b. Note that due to the larger total air-gap, the total effective permeance P in FIG. 9b (and hence corresponding inductance) is still further reduced from permenaces of the separate cores in FIG. 8c and FIG. 8d. 
The main advantage of the coupled-inductor structure is that it can reduce the ripple current on the output side dramatically and even produce zero output ripple current, as first disclosed in U.S. Pat. No. 4,184,197. As disclosed in U.S. Pat. No. 5,790,005 xe2x80x9cLow Profile Coupled Inductors and Integrated Magneticsxe2x80x9d, the inventors E. Santi and S. Cuk have shown that the air-gap position plays the key role in zero ripple current adjustment. When the air-gap is solely placed on the side of input inductor as in FIG. 10a, the total leakage inductance LL effectively appears solely on the output inductor side as in the model of FIG. 10b. Since the converter of FIG. 7a generates identical AC voltages on the input and output inductors, the net AC voltage across this leakage inductance is zero (xcex94v=vL1xe2x88x92vL2=0) leading to zero ripple current (xcex94i2=0) in the output inductor.
Note that the ripple current on the input inductor remains relatively large due to presence of the air-gap. The only way to reduce that ripple would be to reduce the air-gap. Thus, one might be tempted to connect on purpose the coupled-inductors of FIG. 9a into the converter of FIG. 7a so that the output inductor dot-marked end is reversed and connected as in FIG. 11a to the junction between diode CR1 and capacitor C1. Note that with such connection the output inductor DC current I2 will flow out of the dot-marked end. Hence, at least for one duty ratio D=0.5, and provided equal number of turns are used on both windings, a complete DC flux cancellation could be accomplished in the coupled-inductors magnetic core. Thus, the air-gap could be eliminated since the DC-ampere-turns of the two windings cancel. However, elimination of the ripple current is not possible even for this single operating point, since the model in FIG. 11b clearly points out that the small residual leakage inductor would now be subject to an AC voltage, which is two times larger than the input inductor AC voltage vL1 resulting in huge circulating ripple current for both input and output inductors.
Clearly, what is needed is a special switching converter which inherently has the opposing flow of the DC currents in the input and output inductor windings (into the dot-marked end and out of dot-marked end respectively) and yet the respective AC voltage waveforms at the two inductors windings should be in phase with each other. Further constraint is to have identical or closely matching magnitudes of both AC voltages and DC currents. Yet an additional constraint is to maintain the above relationship over a wide operating range, that is a wide change of the duty ratio D. Note that even the first constraint of opposing DC current flows (for the net DC-ampere-turn reduction, if not complete cancellation) and the in-phase waveforms of the respective AC voltages is not realized in the converter of FIG. 7a as well as in all other Coupled-inductors and Integrated Magnetics structures proposed in the past.
Out of a large number of possible switching converters, with input and output inductors, only a handful of them even meet the pre-requisite for coupling them on a common magnetic core, that is to have identical voltage waveforms. Thus, imposing the additional even more severe constraints, such as opposing DC current flows as well as their matching magnitudes, may appear at first too restrictive and impossible to achieve at all. This, however, is not the case, as this invention will demonstrate.
Soft-Switching Advantages and Drawbacks
Another critical performance characteristic of the switching converters is how well switching losses are reduced, since the drive toward smaller size converters has inevitably pushed switching frequencies to very high levels, such as 100 kHz and beyond, even up to 1 MHz. The resulting increase of the switching losses as well as the generated Electromagnetic Interference (EMI) noise have led to the invention of soft switching methods to keep both of these deficiencies under control. One of the prior art methods which provides soft-switching in basic DC-to-DC converters is explained by C. Henze, H. C. Martin and D. W. Parsley, in xe2x80x9cZero-Voltage Switching in High-Frequency Power Converters Using Pulse-Width Modulationxe2x80x9d, IEEE Applied Power Electronics Conference, (IEEE Publication 88CH2504-9) pp33-40, 1988 record, using the buck converter as an example.
In order to obtain lossless zero-voltage switching at a constant switching frequency, the common transistor-diode pair is replaced with composite, current bidirectional switches S and Sxe2x80x2 (when one switch is ON the other is OFF and vice versa) realized in practice with MOSFET transistors. MOSFET switches include an anti-parallel xe2x80x9cbodyxe2x80x9d diode and a parasitic drain-to-source capacitor, thus they can be modeled as ideal switches S and Sxe2x80x2 with a diode and capacitor in parallel as in FIGS. 12a-d. The total switching cycle is broken down into 4 intervals by proper drive timing of the two switches S and Sxe2x80x2 as shown in FIG. 13. Note that with two controllable switches, two well defined transition intervals are introduced during which both switches are OFF. The first transition interval (tN in FIG. 13), starting when switch S is turning OFF (as in FIG. 12a) is also known as the xe2x80x9cnaturalxe2x80x9d transition (DTS to Dxe2x80x2TS transition, or simply D to Dxe2x80x2 transition, where Dxe2x80x2=1xe2x88x92D), since just by turning OFF the switch S, the naturally positive inductor current (represented by the current source on FIG. 12a) charges the parasitic capacitor CS of switch S and discharges parasitic capacitor Cxe2x80x2S of switch Sxe2x80x2 until capacitor Cxe2x80x2S is fully discharged at which time the body diode of switch Sxe2x80x2 clamps the voltage at zero and prevents reverse charging of capacitor Cxe2x80x2S of switch Sxe2x80x2. At that instance, the switch Sxe2x80x2 can be turned ON with zero switching losses (FIG. 12b), since the charge of Cxe2x80x2S was already relocated to capacitance CS of the switch S (charged to Vg). Now in order to perform the reverse process during the Dxe2x80x2 to D transition, the reversal, that is a negative inductor current is needed. The simplest method to accomplish this is to design the output inductor to have a large ripple current, such that its peak-to-peak ripple current is at least 2.5 to 3 times the maximum DC load current. As seen in the inductor current waveform in FIG. 13, the instantaneous inductor current will at some point reverse direction (see FIG. 12c) and look like a negative current source with magnitude IN. Just before the end of complementary interval Dxe2x80x2 the switch Sxe2x80x2 is turned OFF initiating the so called xe2x80x9cforcedxe2x80x9d transition (since the inductor current was intentionally forced by the circuit design to become negative). During this forced transition interval (tF in FIG. 13) the opposite occurs: this negative inductor current charges capacitor Cxe2x80x2S of switch Sxe2x80x2 and discharges capacitor CS of switch S until its voltage VS reaches zero. At that instant body diode clamps the voltage on switch S to zero making it possible for switch S to turn ON at zero voltage in a lossless manner. Hence recycling of the charge stored in the parasitic capacitors CS and Cxe2x80x2S is provided instead of being dissipated each cycle as in xe2x80x9chard-switchingxe2x80x9d.
Even though loss-less switching can be achieved in this very simple manner, and the voltage stresses on the switches are the same as in a basic PWM converter without soft switching, the big disadvantage is that the magnitude of the output inductor ripple current must be at least more than two times greater than the maximum DC load current in order to achieve the soft switching for all operating conditions. Clearly, this soft switching method suffers from the need to create a large ripple current so that a negative instantaneous inductor current is obtained before the end of Dxe2x80x2TS interval in order to accomplish the forced transition. This, in turn, increases the conduction losses and thus diminishes the savings obtained by reduced switching losses. In addition, an increased output capacitor size is needed to absorb this large ripple current and to reduce the output ripple voltage.
An alternative soft switching method which eliminates the need for a large inductor ripple current was proposed by A. Pietkiewicz, S. Cuk, and M. Brkovic in a U.S. Patent No. 5,539,630 xe2x80x9cSoft-Switching Converter DC-to-DC Isolated with Voltage Bi-directional Switches on the Secondary Side of an Isolation Transformerxe2x80x9d for bridge type converters. In their soft switching half-bridge converter, the primary, high voltage switching transition is aided by the DC load current reflected to the primary and hence does not need any ripple current to implement soft switching on the primary side. This method, however, requires two voltage bi-directional switches on the secondary side (each implemented by a series connection of a MOSFET transistor and a diode), which are, due to higher voltage drops and excessive conduction losses, not well suited for low output DC voltage applications.
A large number of various resonant converters and their derivatives, such as quasi-resonant and multi-resonant converters, have been proposed in the prior art. A resonant converter is a power converter in which one or more switching waveforms (either switch-voltage or switch-current) is distorted into sinusoidal ringing waveform, with either zero-voltage or zero-current crossing, which enables a corresponding zero-voltage or zero-current switching, thereby reducing switching losses. Even though these resonant converters are effective in reducing switching losses, the very nature of their operation substantially increases either RMS currents or voltage stresses on the devices, and hence ultimately increases conduction losses thereby diminishing savings due to reduced switching losses.
From the above review, it is clear that a new soft switching method is needed which reduces the switching losses without introducing all the other undesirable features associated with prior art soft switching methods and thereby preserves the high overall efficiency. This invention introduces such a novel switching converter, which in its basic operation has the inherent capability of zero-voltage soft switching.
Although much progress has been made in the past, two fundamental problems needed to be solved before further substantial improvements in efficiency and size reductions could be made:
1. Presently, switching DC-to-DC converters utilize magnetic components with a large air-gap in order to avoid saturation due to DC currents in their windings and the presence of DC flux. Large loss of inductance is compensated either by an increase of the switching frequency or by an increase of magnetic core size, or both, with consequent direct reduction of efficiency and increase of the magnetic core and consequently the converter size and weight. A converter with a new magnetic circuit is needed that will eliminate DC flux in the core and thus enable magnetics to be built on a ferromagnetic core without any air-gap and without any wasteful DC energy storage. In that case, the ferromagnetic material will be fully utilized in its ability to generate large inductances and effectively provide filtering, even with small size magnetics and at moderate switching frequencies.
2. A number of soft switching methods proposed inthe past, while producing beneficial zero voltage switching and reduction of switching losses, typically suffer from increase of conduction losses, or significantly higher voltage or current stresses on the devices compared to their Pulse Width Modulation (PWM) drive, thus ultimately resulting in diminished savings. Therefore, soft switching methods are needed without such detrimental side loss mechanisms.
Novel DC Transformer switching converter and Isolated DC Transformer switching converter described below successfully solve both of the above problems.
A primary objective of this invention is to provide a switching DC-to-DC converter that, through the use of a new magnetic device, namely a DC Transformer, achieves simultaneously high efficiency, high overload capability, small size and weight, low input and output ripple currents and low EMI noise.
Another objective is to provide a switching DC-to-DC converter using a special magnetic device, a DC Transformer, which combines all magnetic components of the converter onto a single magnetic core to enable zero total DC-ampere-turns, zero DC flux and hence no air-gap in the magnetic core. Conventional separate inductors or Coupled-inductors and Integrated Magnetic structures have large DC flux and thus need to include a large air-gap to prevent saturation of the magnetic core with consequent large loss of inductance and corresponding performance degradation. This DC Transformer without an air-gap in its magnetic path, has high DC output current overload capability, small size and weight, and provides desirable ripple-free DC input and DC load currents. The DC stored energy is also reduced to zero leading to corresponding increase in efficiency.
Yet another objective is to provide further increase in efficiency and reduction of the size of a DC-to-DC switching converter through the use of two kinds of soft-switching operation: a simpler one, designated partial soft switching, with only timing adjustments of the drives for four MOSFET-like semiconductor switches providing substantial switching loss reduction, and more complex one that includes a resonant inductor already present as a leakage inductance of an isolation transformer to further improve efficiency.
These and other objectives are achieved in a DC Transformer switching converter having input, middle and output inductor windings placed on a common magnetic core to form an effective non-isolated DC Transformer.
The standard AC voltage test is then performed to determine the ends of the DC Transformer windings at which AC voltages are in phase, and these ends, designated as dot-marked ends, are connected as follows: input inductor dot-marked end to the input DC source terminal, output inductor dot-marked end to the output DC load terminal, and the middle inductor dot-marked end to the common input terminal and common output terminal. An input capacitor is connected between unmarked ends of the input inductor and the middle inductor. An input switch periodically connects the unmarked end of input inductor to the common input terminal, and operates in phase with an output switch which connects the unmarked end of output inductor to the common output terminal, i.e., both switches are ON for an interval DTS and OFF for a complementary switching interval Dxe2x80x2TS=(1xe2x88x92D)TS. An output complementary switch periodically connects the unmarked end of the output inductor to the unmarked end of the middle inductor. A branch comprising the complementary input switch (operating in-phase with the complementary output switch) and an auxiliary capacitor in series is then connected to the converter so that the current through the auxiliary capacitor during complementary interval Dxe2x80x2TS is equal to the sum of the input inductor current and the middle inductor current reduced by the output inductor current, with input inductor and middle inductor currents flowing into their dot-marked ends and output inductor current flowing out of its dot-marked end. The implementation of identical number of turns for all three windings will insure zero total DC-ampere-turns and insure that the DC Transformer, which has no air-gap in its magnetic flux path, will provide high DC overload capability. The foregoing precise connections of the DC Transformer to the remaining switching converter circuitry is necessary for full DC Transformer performance.
Galvanic isolation between the source and the load is required in many practical applications. This is accomplished in another embodiment of the present invention by replacing the middle inductor with an isolation transformer, which provides both galvanic isolation as well as an additional voltage-scaling factor of the output DC voltage equal to the ratio of turns of isolation transformer secondary to its primary number of turns. This Isolated DC Transformer retains all of the properties of its non-isolated DC Transformer counterpart, provided the isolation transformer windings retain the dot marking polarity of the middle inductor winding and provided that the number of turns are chosen as follows: input inductor number of turns equal to transformer primary number of turns, and output inductor number of turns equal to transformer secondary number of turns. This will insure that the total DC ampere-turns are zero and that a new magnetic component, the Isolated DC Transformer which has no air-gap in its magnetic flux path, provides the high DC overload capability. The foregoing precise connection of the Isolated DC Transformer to the remaining switching converter structure is necessary, for full performance of the Isolated DC Transformer.
In yet another embodiment of the present invention, a non-isolated DC Transformer switching converter uses only the drive timing of four MOSFET-like switching devices to accomplish partial soft switching to substantially reduce the switching losses of the input high voltage switches which dominate the switching losses. The drive timing is so adjusted as to provide two transition intervals in each switching period TS, during which both an input switch and a complementary input switch are OFF. The first transition interval is initiated when the input switch is turned OFF. When the voltage of the complementary input switch reduces to zero that switch is turned ON with ideally zero switching losses. The second transition interval is initiated when the complementary input switch is turned OFF. When the voltage of the input switch is substantially reduced, it is turned ON with reduced switching losses and a partial soft switching cycle is completed.
In still another embodiment of the present invention, the Isolated DC Transformer switching converter relies on the leakage inductance of the isolation transformer to assist and accomplish full soft switching. The drive timing is so adjusted as to provide two transition intervals in each switching period TS, during which both an input switch and a complementary input switch are OFF. The first transition interval is initiated by turning-OFF of the input switch. When the voltage of the complementary input switch reduces to zero, the complementary input switch is turned ON with ideally zero switching losses. The second transition interval is initiated when the complementary input switch is turned OFF and the output switch is simultaneously turned ON to force the resonant discharge of a parasitic capacitor across the input switch. When it is fully discharged, the input switch is turned ON at zero voltage to result in zero switching losses.
The novel features that are considered characteristic of this invention are set forth with particularity in the appended claims. The invention will best be understood from the following description when read in connection with the accompanying drawings.